The present invention relates to a switching power supply circuit which may be used as a power supply for electronic apparatus.
A switching power supply circuit which includes a switching converter of a voltage resonance type is a soft switching power supply circuit. In such circuit, a switching output pulse voltage and switching output current produced by the switching converter and supplied to an insulating converter transformer may have smooth waveforms. As a result, the switching converter may generate relatively low noise. Furthermore, such switching converter may be formed from a relatively small number of parts.
FIG. 11 illustrates a switching power supply circuit of the voltage resonance type. Such switching power supply circuit is operable with a commercial ac power supply AC of 100 V which may be available in Japan or the United States and is usable with a maximum load power of 150 W or more.
The switching power supply circuit shown in FIG. 11 includes a rectifier smoothing circuit for rectifying and smoothing the commercial ac power supply AC. The rectifier smoothing circuit is formed as a voltage multiplying rectifier circuit composed of a pair of rectifier diodes Di1 and Di2 and a pair of smoothing capacitors Ci1 and Ci2. The voltage multiplying rectifier circuit may produce a dc input voltage equal to approximately twice a dc input voltage Ei which is equal to a peak value of the ac input voltage VAC. For example, if the ac input voltage VAC is 144 V, then the dc input voltage 2Ei is approximately 400 V
The voltage multiplying rectifier circuit is adopted as the rectifier smoothing circuit so as to enable a relatively heavy load from the ac input voltage of 100 V and the maximum load power of 150 W or more. In other words, the dc input voltage is set to twice the normal voltage to suppress the amount of inflow current to a switching converter in the next stage so as to improve the reliability of the components of the switching power supply circuit.
An inrush current limiting resistor Ri is inserted in a rectifier current path of the voltage multiplying rectifier circuit shown in FIG. 11. As a result, inrush current which may flow into the smoothin capacitors during the initial supply of power may be suppressed.
The switching power supply circuit of FIG. 11 may include a switching converter of the voltage resonance type having a self-excited construction and including a single switching element Q1. Such switching element may be a high voltage withstanding bipolar transistor (BJT: junction transistor). The collector of the switching element Q1 is connected to an end of a primary winding N1 of an insulating converter power isolation transformer (PIT), and the emitter of the switching element Q1 is grounded. The base of the switching element Q1 is coupled to the positive electrode side of the smoothing capacitor Ci2 (rectified smoothed voltage Ei) through a starting resistor RS. As a result, during a starting operation, the current supplied to the base of the switching element Q1 may be rectified and smoothed. Further, a resonance circuit for self-excited oscillation is connected between the base of the switching element Q1 and the primary side ground and is formed from a series connection of an inductor LB, a resonance capacitor CB, a detection driving winding NB, and a damping resistor RB. The detection driving winding NB is wound on the insulating converter PIT and together with the inductor LB provides the inductance for setting a switching frequency.
A clamp diode DD is arranged between the base of the switching element Q1 and the primary side ground and forms a path for damper current which flows when the switching element Q1 is off.
A resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. Based on the capacitance of the resonance capacitor Cr and a combined inductance (L1 and LR) obtained from a series connection of the primary winding N1 of the insulating converter transformer PIT and a controlled winding NR of an orthogonal control power regulating transformer (PRT), the resonance capacitor Cr forms a resonance circuit of a voltage resonance type converter. When the switching element Q1 is off, a voltage resonance type operation may be obtained by the resonance circuit which causes the voltage Vcr across the resonance capacitor Cr to exhibit a pulse wave of a sine waveform.
One end of the primary winding N1 of the PIT is connected to the collector of the switching element Q1, and the other end of the primary winding N1 is connected to the controlled winding NR of the PRT.
The PIT transmits a switching output of the switching element Q1 to the secondary side.
On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 appears in the secondary winding N2. A secondary side parallel resonance capacitor C2 is connected in parallel to the secondary winding N2 so as to form a parallel resonance circuit. The alternating voltage induced in the secondary winding N2 is converted into a resonance voltage by the parallel resonance circuit. Such resonance voltage is supplied to two half-wave rectifier circuits in which one such half-wave rectifier circuit includes a rectifier diode D01 and a smoothing capacitor C01 and the other half-wave rectifier circuit includes a rectifier diode D02 and a smoothing capacitor C02. The two half-wave rectifier circuits produce two different dc output voltages E01 and E02. The rectifier diodes D01 and D02 may be high-speed type rectifier diodes so as to rectify the alternating voltage of a switching period.
The control circuit 1 is an error amplifier which may compare a dc output voltage of the secondary side with a reference voltage and supply a dc current corresponding to an error therebetween as a control current to the control winding NC of the orthogonal control transformer PRT. Here, the dc output voltage E01 and the dc output voltage E02 may be supplied to the control circuit 1 as a detection voltage and as an operation power supply, respectively.
As an example, if the dc output voltage E02 of the secondary side varies in response to a variation of the ac input voltage VAC or the load power, then the control current which is to flow through the control winding NC may be varied within the range of 10 mA to 40 mA by the control circuit 1. As a result, the inductance LR of the controlled winding NR may vary within the range of 0.1 mH to 0.6 mH.
Since the controlled winding NR forms a resonance circuit which may perform a voltage resonance type switching operation as previously described, the resonance condition of the resonance circuit may vary with respect to the switching frequency which is fixed. Across the parallel circuit of the switching element Q1 and the resonance capacitor Cr, a resonance pulse of a sine waveform may appear due to the resonance circuit corresponding to an off period of the switching element Q1 and the width of the resonance pulse may be variably controlled by the variation of the resonance condition of the parallel resonance circuit. As such, a pulse width modulation (PWM) control operation for a resonance pulse may be obtained. The PWM control of the resonance pulse width may occur during the off period of the switching element Q1 and, as a result, the on period of the switching element Q1 is variably controlled in the condition wherein the switching frequency is fixed. Since the on period of the switching element Q1 is variably controlled in this manner, the switching output transmitted from the primary winding N1 (which forms the parallel resonance circuit to the secondary side) varies, and the level or levels of the dc output voltages E01 and E02 of the secondary side vary. Consequently, the secondary side dc output voltage E01 or E02 is controlled to a constant voltage. Such constant voltage control method is hereinafter referred to as an inductance control method.
FIG. 12 illustrates another switching power supply circuit of the voltage resonance type. Elements in FIG. 12 similar to those in FIG. 11 are denoted by the same reference characters and, in the interest of brevity, a further description thereof is omitted herein.
In the power supply circuit of FIG. 12, a controlled winding of an orthogonal control transformer PRT is provided on the secondary side. Such controlled winding of the orthogonal control transformer PRT may include two controlled windings NR and NR1. The controlled winding NR is arranged in series between an end of the secondary winding N2 and the anode of the rectifier diode D01. The controlled winding NR1 is arranged in series between a tap output of the secondary winding N2 and the anode of the rectifier diode D02. In such configuration, a parallel resonance circuit of the secondary side is formed which includes inductance components of the controlled windings NR and NR1.
In the arrangement of FIG. 12 wherein the controlled windings (NR and NR1) of the orthogonal control transformer PRT are provided on the secondary side, the orthogonal control transformer PRT operates such that, as the inductance of the controlled winding NR is varied in accordance with an inductance control method, the pulse width of a resonance voltage V2 of the secondary side parallel resonance capacitor C2, that is, the continuity angle of the secondary side rectifier diodes, is variably controlled. Such control of the output level on the secondary side enables constant voltage control to be achieved.
The insulating converter transformer PIT provided in the power supply circuits of FIGS. 11 and 12 is illustrated in FIG. 13. As shown therein, the insulating converter transformer PIT includes an EE-shaped core having a pair of E-shaped cores CR1 and CR2 which may be made of a ferrite material. These E-shaped cores may be combined to each other such that magnetic legs thereof are opposed to each other and such that a gap is not provided between the middle magnetic legs. The primary winding N1 and the secondary winding N2 are wound separately from each other on the central magnetic legs of the EE-shaped core using a bobbin B. As a result, a loose coupling (for example, a coupling coefficient k may have a value of approximately 0.9) may be obtained between the primary winding N1 and the secondary winding N2.
In the insulating converter transformer PIT, a mutual inductance M between inductance L1 of the primary winding N1 and inductance L2 of the secondary winding N2 may have a value +M (additive mode) or a value xe2x88x92M (subtractive mode) depending upon the relationship between the polarities (winding directions) of the primary winding N1 and the secondary winding N2 and the connection of the rectifier diodes D01 and D02. For example, if such components have a configuration as shown in FIG. 14A, then the mutual inductance is +M; however, if such components have a configuration as shown in FIG. 14B, then the mutual inductance is xe2x88x92M.
FIGS. 15A to 15C illustrate operation waveforms in a switching period of the power supply circuit of FIG. 11. In these figures, reference characters TON and TOFF denote periods wherein the switching element Q1 is on and off, respectively, and reference characters DON and DOFF denote periods wherein the rectifier diode D01 on the secondary side is on and off, respectively.
The resonance voltage Vcr across the switching element Q1 and resonance capacitor Cr has a waveform similar to a pulse of a sine waveform within a period TOFF (as shown in FIG. 15A) wherein the switching element Q1 is off and the operation of the switching converter is a voltage resonance type operation. The peak level of the pulse of the resonance voltage Vcr is approximately 1,800 V which is due to the impedance of the resonance circuit of the primary side of the voltage resonance converter acting upon the dc input voltage of 2Ei obtained by the voltage multiplying rectification.
With regard to the operation of the secondary side, the rectifier diode D01 operates such that rectified current flows within a period DON which is approximately equal to the period TON of the switching element Q1 as shown in FIG. 15C. This operation is based on the +M (additive mode) mutual inductance described above with reference to FIG. 14. A substantially similar operation timing is also obtained with regard to the rectifier diode D02.
As a result of the above described rectification operation, the resonance voltage V2 across the secondary side parallel resonance capacitor C2 becomes a sine waveform having a peak level equal to twice to 3.5 times the dc output voltage E0 (E01 or E02) within the period DOFF wherein the rectifier diode D01 is off, and a voltage level equal to the dc output voltage E0 (E01 or E02) within the period DON wherein the rectifier diode D01 is on, as shown in FIG. 15B.
In the voltage resonance converters described above with reference to FIGS. 11 to 15C, a dc input voltage having a level of 2Ei is obtained using the voltage multiplying rectification system so as to satisfy the condition of an ac input voltage VAC of AC 100 V and a maximum load power of 150 W or more. Therefore, as described hereinabove with reference to FIG. 15A, the resonance voltage Vcr of 1,800 V appears across the switching element Q1 and the parallel resonance capacitor Cr when the switching element Q1 is off.
Therefore, the switching element Q1 and the resonance capacitor Cr should be able to withstand a high voltage. As a result, the switching element Q1 and the resonance capacitor Cr have relatively large sizes. Furthermore, and particularly when a high withstanding voltage switching element Q1 is used, since such element is relatively high in saturation voltage VCE (SAT) and long in storage time tSTG and fall time tf and is relatively low in current amplification factor hFE, it may be difficult to set the switching frequency to a relatively high value. A low value or a decrease of the switching frequency may increase the switching loss and the drive power which may increase the power loss of the power supply circuit.
Further, in the power supply circuits shown in FIGS. 11 and 12, the controlled winding NR of the orthogonal control transformer PRT is connected in series to one of the primary winding N1 and the secondary winding N2. Such arrangement may increase a leakage inductance component of the insulating converter transformer PIT.
As a countermeasure, the entire power supply circuit may be arranged in an aluminum shield case having vent holes formed therein and a connector for connecting input and output lines may be mounted on a circuit board. However, such countermeasure may increase the size and weight of the power supply circuit and may increase the complexity of the fabrication thereof.
It is an object of the present invention to provide a switching power supply circuit which can handle a relatively high power load, has a relatively high power conversion efficiency, and has a relatively small size and light weight.
According to an aspect of the present invention, a switching power supply circuit is provided which comprises a rectifier smoothing circuit for receiving an ac power supply, producing a rectified smoothed voltage having a level equal to that of the ac power supply and outputting the rectified smoothed voltage as a dc input voltage; an insulating converter transformer for transmitting a primary side output to a secondary side, in which the insulating converter transformer has a gap formed therein so that a coupling efficient (k) for a loose coupling is obtained; a switching circuit including a switching element for switching the dc input voltage on and off so as to be outputted to a primary winding of the insulating converter transformer; a primary side resonance circuit formed from a leakage inductance component from the primary winding of the insulating converter transformer and a capacitance of a resonance capacitor to enable the switching circuit to operate as a voltage resonance type; a secondary side parallel resonance circuit including a secondary winding of the insulating converter transformer and a secondary side parallel resonance capacitor connected such that a parallel resonance circuit is formed from a leakage inductance component of the secondary winding of the insulating converter transformer and a capacitance of the secondary side parallel resonance capacitor; a dc output voltage production circuit for receiving an alternating voltage obtained at the secondary winding of the insulating converter transformer and performing a half-wave rectification operation by an additive mode for the alternating voltage to produce a secondary side dc output voltage; and a constant voltage control circuit for varying a switching frequency of the switching element in response to a level of the secondary side dc output voltage to perform constant voltage control of the secondary side output voltage.
In the present switching power supply circuit, the insulating converter transformer has a loose coupling, and the resonance circuit for forming a voltage resonance converter on the primary side and the parallel resonance circuit on the secondary side form a composite resonance converter. Further, the constant voltage control is performed by controlling the switching frequency of the switching element which forms the voltage resonance converter of the primary side. The switching power circuit can thus operate to vary the switching frequency within a high frequency range.
Instead of a voltage multiplying rectifier circuit, the present switching power supply circuit may, on the primary side, include a full-wave rectifier circuit for producing a rectified smoothed voltage equal to the level of the ac input voltage thereto.
Therefore, the present switching power supply circuit may include a composite resonance converter wherein a voltage resonance converter is provided on the primary side and a parallel resonance circuit is provided on the secondary side, and a gap is formed in a middle magnetic leg of an insulating converter transformer so that the insulating converter transformer may have a loose coupling condition and a coupling coefficient higher than a predetermined value and a half-wave rectifier circuit of an additive mode is provided on the secondary side. The switching frequency is varied to perform constant voltage control.
In the present switching power supply circuit, constant voltage control may be performed by switching frequency control. Further, the switching frequency may be set to a relatively high level as compared to other circuits wherein inductance control of the insulating converter transformer is performed while the switching frequency is fixed or the width of a voltage resonance pulse is variably controlled.
When the switching frequency is set to a relatively high level, power loss by switching decreases and, as a result, an increase in power conversion efficiency over a wide range of load conditions can be achieved.
Further, since the parallel resonance circuit on the secondary side operates with constant voltage control, the range of the constant voltage control may be expanded.
During an operation of the switching power supply circuit when the load is relatively heavy, the switching frequency of the switching element may be controlled so as to increase the on period of the switching element. Since relatively high levels of primary side resonance current and secondary side resonance current may be supplied during the period, the switching power supply circuit can handle the heavy load condition. As a result, an increase in the maximum load power can be achieved with the present switching power supply circuit. Accordingly, the present switching power supply circuit may be applied to an apparatus which exhibits a large fluctuation in the load.
Since the switching power supply circuit can increase the maximum load power, it can sufficiently handle the condition described above even if it is constructed such that instead of a voltage multiplying rectifier circuit an ordinary full-wave rectifier circuit is employed on the primary side so that a rectified smoothed voltage corresponding to the ac input voltage level may be inputted.
For a conventional switching power supply circuit to handle the condition described above, it uses a voltage multiplying rectifier circuit to produce a rectified smoothed voltage equal to twice the ac input voltage level. Therefore, in such circuit, the switching element or the resonance capacitor on the primary side should have a voltage withstanding property against a switching voltage generated in response to the rectified smoothed voltage level.
On the other hand, with regard to the present switching power supply circuit, since the primary side resonance voltage which depends upon a rectified smoothed voltage level is much lower than that of the conventional switching power supply circuit as a result of the equal voltage rectifier circuit and the ability to raise the switching frequency to a high level, the switching element or the primary side resonance capacitor may have a voltage withstanding property lower than that of the conventional switching power supply circuit and may have a smaller size, lower weight and superior characteristics compared to that of the conventional switching power supply circuit.
Thus, the present switching power supply circuit (which may include a voltage resonance converter) may have a relatively small size and weight, may provide a relatively high power conversion efficiency and improved characteristics such as a load power characteristic as compared to the conventional switching power supply circuit.
Other objects, features and advantages according to the present invention will become apparent from the following detailed description of illustrated embodiments when read in connection with the accompanying drawings in which corresponding components are identified by the same reference numerals.